Fill in your specification on the left-hand side. You can load some common core and geometry properties.

The current is specificed is used to estimate losses. *I*_{ripple} is the peak-to-peak of the ripple current. *I*_{peak} is the highest current experienced. For a triangular current, these will be the same. For a sinusoidal current, *I*_{ripple} = 2 *I*_{peak}

The inductor design process is as follows: —

- Fill in the specifications, including
*k*_{Cu}as the maximum packing factor. Don't forget to press**Design now**. - Note whether the design is possible. This is shown by the solid black line entering the shaded region in graph 1.
- Use graph 2 to identify the conductor diameter. The shaded region in graph 2 is the valid region from graph 1.
- Choose the nearest suitable conductor diameter and update the specification by changing
*D*. Graph 3 will update. Select a number of turns for low power loss and appropriate*k*_{Cu}. - From graph 1, read the required air-gap length for your number of turns.

#### Processing messages (if any)

## Air gap and turns for correct inductance

For a particular air gap, fewer turns means less saturated and power core loss. It also means reduced inductance.

As you increase the air gap, the number of turns needed to achieve the same flux and power loss in the core increases. The number of turns to achieve the desired inductance also increases, but more slowly. At a large enough air-gap, the desired inductance can be achieved within flux and power loss limits.

The maximum practical air gap is normally taken as a tenth the square root of the core area - any larger and significant fringing occurs.

## Power loss with *k*_{Cu} = unknown

## Power loss with *D* = unknown

This website helps design inductors by providing estimations of the air-gap and losses against limits of saturation and air-gap size. The starting point of the design is the link between core reluctance and inductance.

where \(L\) is the inductance, \(N\) is the number of turns and \(S\) is the combined reluctance of the core and the gap. For a gapped inductor, \(S={A_\textrm L}^{-1} + S_\textrm{gap}\), hence

The gap reluctance is calculated assuming the flux remains constrained to the same area.

The operation of inductors depends on their current waveform and amplitude. For this design tool, we assume an offset sinusoid with formula

This means the RMS current is

This definition is useful as *I*_{peak} and *I*_{ripple} are easy to find for all waveforms, while the RMS, peak and the sinusoidal component are easily analysed.

Magnetic cores saturate above a certain flux density. This maximum flux density places a limit on the number of turns, which varies with air-gap length

Similarly, the maximum core loss can be specified, producing a separate limit.

This is based on the Steinmetz equation. Note that the offset of the sine wave modelled is not taken into account. Core loss, especially for nearly-saturated cores, is likely to be overestimated due to reduced hysteresis.

Estimations for the overall power loss are found by adding the core loss to winding loss. Winding loss can be estimated either accounting for skin depth or not

Finally, the air gap must not be so wide that significant fringing occurs. Fringing can cause proximity effect (and increased losses) and also causes reluctance to be overestimated. For this design tool, the air gap is limited to